Radar device

ABSTRACT

A radar device includes a radar transmitting circuit that transmits radar signals from a transmission array antenna, and a radar receiving circuit that receives returning wave signals, where the radar signals have been reflected at a target, from a receiving array antenna. One of the transmitting array antenna and the receiving array antenna includes multiple first antennas of which phase centers are laid out along a first axis direction. The other of the transmitting array antenna and the receiving array antenna includes multiple second antennas of which phase centers are laid out at a second spacing along a second axis direction that is different from the first axis direction. The multiple first antennas include multiple antennas of which the phase centers are laid out at a first spacing, and multiple antennas of which the phase centers are laid out at a third spacing that is different from the first spacing.

BACKGROUND 1. Technical Field

The present disclosure relates to a radar device.

2. Description of the Related Art

In recent years, radar devices are being studied that use shortwavelength radar transmission signals, including microwaves ormillimeter waves that yield high resolution. There also is demand fordevelopment of radar devices that detect, in addition to vehicles,objects (targets) including pedestrians, over a wide angle range(wide-angle radar devices), in order to improve safety outdoors.

There also has been proposed a radar device having a configuration thathas multiple antennas (antenna array) at a transmitting branch inaddition to a receiving branch, and that performs beam scanning bysignal processing using transmitting/receiving array antennas (alsoreferred to as Multiple Input and Multiple Output (MIMO) radar) (e.g.,see P. P. Vaidyanathan, P. Pal, Chun-Yang Chen, “MIMO radar withbroadband waveforms: Smearing filter banks and 2D virtual arrays, “IEEEAsilomar Conference on Signals, Systems and Computers, pp. 188-192, 2008(hereinafter “VAIDYANATHAN et al”).

A MIMO radar enables configuration of virtual receiving array antennas(hereinafter referred to as virtual receiving array) equivalent to theproduct of the transmitting antenna element count and receiving antennaelement count as a maximum, by innovative layout of antenna elements ina transmitting/receiving array antenna. This is advantageous in that theeffective aperture length of the array antenna can be increased with asmall number of elements.

Also, MIMO radar can be applied in cases of performing three-dimensionalmeasurement by beam scanning in the two dimensions of the verticaldirection and horizontal direction, besides one-dimensional scanning inthe vertical direction or the horizontal direction (e.g., see JapaneseUnexamined Patent Application Publication (Translation of PCTApplication) No. 2017-534881 and VAIDYANATHAN et al).

SUMMARY

One non-limiting and exemplary embodiment provides a radar devicecapable of three-dimensional measurement, while suppressing sidelobesand securing high resolution in the horizontal direction.

In one general aspect, the techniques disclosed here feature a radardevice including a radar transmitting circuit that transmits radarsignals from a transmission array antenna, and a radar receiving circuitthat receives returning wave signals, where the radar signals have beenreflected at a target, from a receiving array antenna. One of thetransmitting array antenna and the receiving array antenna includes aplurality of first antennas of which phase centers are laid out along afirst axis direction. The other of the transmitting array antenna andthe receiving array antenna includes a plurality of second antennas ofwhich phase centers are laid out at a second spacing along a second axisdirection that is different from the first axis direction. The pluralityof first antennas include a plurality of antennas of which the phasecenters are laid out at a first spacing, and a plurality of antennas ofwhich the phase centers are laid out at a third spacing that isdifferent from the first spacing.

According to an aspect of the present disclosure, a radar device capableof three-dimensional measurement, while suppressing sidelobes andsecuring high resolution in the horizontal direction, can be provided.

It should be noted that general or specific embodiments may beimplemented as a system, a device, a method, an integrated circuit, acomputer program, or a recording medium, or may be realized by anycombination of system, device, method, integrated circuit, computerprogram, and recording medium.

Additional benefits and advantages of the disclosed embodiments willbecome apparent from the specification and drawings. The benefits and/oradvantages may be individually obtained by the various embodiments andfeatures of the specification and drawings, which need not all beprovided in order to obtain one or more of such benefits and/oradvantages.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a bock diagram illustrating an example of the configuration ofa radar device according to a first embodiment;

FIG. 2 is a block diagram illustrating an example of the configurationof a radar transmitting unit according to the first embodiment;

FIG. 3 is a block diagram illustrating an example of radar transmissionsignals according to the first embodiment;

FIG. 4 is a diagram illustrating an example of time division switchingoperations of a transmitting antenna by a control unit according to thefirst embodiment;

FIG. 5 is a block diagram illustrating an example of anotherconfiguration of a radar transmission signal generating unit accordingto the first embodiment;

FIG. 6 is a block diagram illustrating an example of the configurationof a radar receiving unit according to the first embodiment;

FIG. 7 is a diagram illustrating an example of transmission timing ofradar transmission signals and measurement range of the radar deviceaccording to the first embodiment;

FIG. 8 is a diagram illustrating a three-dimensional coordinate systemused to describe operations of a direction estimating unit according tothe first embodiment;

FIG. 9 is a diagram illustrating a first layout example of receivingantennas in a receiving array antenna according to the first embodiment;

FIG. 10A is a diagram illustrating the first layout example of receivingantennas in a receiving array antenna according to the first embodiment;

FIG. 10B is a diagram illustrating the first layout example oftransmitting antennas in a transmitting array antenna according to thefirst embodiment;

FIG. 10C is a diagram illustrating the layout of a virtual receivingarray according to the first layout example;

FIG. 11A is a diagram illustrating the layout of receiving antennas in areceiving array antenna according to a first comparative example;

FIG. 11B is a diagram illustrating the layout of transmitting antennasin a transmitting array antenna according to the first comparativeexample;

FIG. 11C is a diagram illustrating the layout of a virtual receivingarray according to the first comparative example;

FIG. 12 is a cross sectional view of two-dimensional beams according tothe first layout example and the first comparative example at 0 degreesin the second axis direction, taken along the first axis direction;

FIG. 13 is a diagram illustrating the layout of a virtual receivingarray according to a second layout example;

FIG. 14A is a diagram illustrating a second layout example of receivingantennas in a receiving array antenna according to the first embodiment;

FIG. 14B is a diagram illustrating the layout of a virtual receivingarray according to the second layout example;

FIG. 15 is a cross sectional view of two-dimensional beams according tothe first layout example and the second layout example at 0 degrees inthe second axis direction, taken along the first axis direction;

FIG. 16A is a diagram illustrating a third layout example oftransmitting antennas in a transmitting array antenna according to asecond embodiment;

FIG. 16B is a diagram illustrating the layout of a virtual receivingarray according to the third layout example;

FIG. 17 is a diagram illustrating the layout of a virtual receivingarray correlation vector according to a third comparative example;

FIG. 18 is a cross sectional view of two-dimensional beams according tothe third layout example and the third comparative example at 0 degreesin the second axis direction, taken along the first axis direction;

FIG. 19 is a diagram illustrating a fourth layout example oftransmitting antennas in a transmitting array antenna according to thesecond embodiment;

FIG. 20 is a block diagram illustrating an example of the configurationof a radar receiving unit according to a third embodiment;

FIG. 21 is a diagram illustrating the fourth layout example oftransmitting antennas in a transmitting array antenna according to thethird embodiment;

FIG. 22 is a diagram illustrating an example of the layout of antennaelements in a transmitting array antenna according to the fourth layoutexample; and

FIG. 23 is a diagram illustrating an example of time division switchingcontrol of a first antenna group and a second antenna group according tothe third embodiment.

DETAILED DESCRIPTION

A type of radar device called a pulse radar device, which repeatedlyemits pulse waves for example, is known. Reception signals of awide-angle pulse radar that detects vehicles and pedestrians over a widerange are signals where multiple reflected waves from nearby targets(e.g., vehicles) and distant targets (e.g., pedestrians) areintermingled. Accordingly, (1) a configuration where the radartransmitting unit transmits pulse waves or pulse modulation waves havingautocorrelation properties of low-range sidelobes (hereinafter referredto as low-range sidelobe properties) is being studied, and (2) aconfiguration where the radar transmitting unit has a broad receptiondynamic range is being studied.

Wide-angle radar device configurations include the following two. Afirst configuration transmits radar waves by mechanically orelectronically scanning pulse waves or modulation waves using anarrow-angle (beam width of around several degrees) directional beam,and receives reflected waves using a narrow-angle directional beam.Tracking capabilities of fast-moving targets deteriorates with thisconfiguration, since the number of times of scanning to obtain highresolution increases.

A second configuration receives reflected waves by an array antenna madeup of multiple antennas (multiple antenna elements) at the receivingbranch, and uses a technique of estimating the arrival angle ofreflected waves (direction of arrival (DOA) estimation) by a signalprocessing algorithm based on reception phase difference as to antennaelement spacing. This configuration enables reduction of the scanningtime since the arrival angle can be estimated at the receiving brancheven if scanning spacings of transmission beams at the transmittingbranch are thinned out, and accordingly tracking capabilities are higheras compared to the first configuration. Examples of DOA estimationtechniques include Fourier transform based on matrix operations, Caponand linear prediction (LP) based on inverse matrix operations, andMUltiple SIgnal Classification (MUSIC) and estimation of signalparameters via rotational invariance techniques (ESPRIT) based oneigenvalue operations.

A MIMO radar that performs beam scanning using multiple antenna elementsat the transmitting branch in addition to the receiving branch transmitssignals multiplexed using time division, frequency division, or codedivision, from multiple transmitting antenna elements. Signals reflectedat surrounding objects are received by multiple receiving antennaelements, and the multiplexed transmission signals are separated fromeach of the received signals, and thus received.

Further, a MIMO radar enables configuration of virtual receiving arrayantennas (virtual receiving array) equivalent to the product of thetransmitting antenna element count and receiving antenna element countas a maximum, by innovative layout of antenna elements in atransmitting/receiving array antenna. Accordingly, propagation channelresponse indicated by the product of the transmitting antenna elementcount and receiving antenna element count can be obtained. Also, theeffective aperture length of the array antenna can be virtuallybroadened with a small number of elements, and angular resolution can beimproved by appropriately laying out spacings betweentransmitting/receiving antenna elements.

Antenna element configurations in a MIMO radar are generally classifiedinto configurations using a single antenna element (hereinafter referredto as single antenna) and configurations using a sub-array arrangementof multiple antenna elements (hereinafter referred to as sub-array). Ina case of using a single antenna, properties are exhibited with broaddirectionality as compared to a case of using a sub-array, but antennagain is relatively lower. Accordingly, in order to improve the receptionsignal to noise ratio (SNR) of the returning wave signals, either moreaddition processing has to be performed in the reception signalprocessing, or the antenna has to be configured using multiple singleantennas, for example.

On the other hand, in a case of using a sub-array, multiple antennaelements are included in one sub-array, unlike the case of using asingle antenna, so the physical size of the antenna increases andantenna gain in the main beam direction can be increased. Specifically,the physical size of the sub-array is around or greater than thewavelength of the radio frequency (carrier frequency) of thetransmission signals.

Also, MIMO radar can be applied in cases of performing two-dimensionalbeam scanning in the vertical direction and horizontal direction,besides one-dimensional scanning in the vertical direction or thehorizontal direction (e.g., see Japanese Unexamined Patent ApplicationPublication (Translation of POT Application) No. 2017-534881 andVAIDYANATHAN et al). For example, there is demand for a MIMO radarcapable of long-distance two-dimensional beam scanning that is used foronboard purposes and so forth to have angular estimation capabilities inthe vertical direction, in addition to high resolution in the horizontaldirection on the same level as a MIMO radar that performsone-dimensional beam scanning in the horizontal direction.

However, there are cases where the number of antenna elements of thetransmitting and receiving branches is restricted, to reduce the sizeand cost of the MIMO radar. For example, in a case where there is arestriction of around four transmitting antenna elements and around fourreceiving antenna elements, the aperture length in the verticaldirection and horizontal direction is restricted in the planar virtualreceiving array of the MIMO radar. Restricting the aperture lengthresults in lower resolution in the vertical direction and horizontaldirection.

For example, there is demand for a MIMO radar capable of long-distancetwo-dimensional beam scanning used for onboard purposes and so forth tohave angular estimation capabilities in the vertical direction, inaddition to high resolution in the horizontal direction on the samelevel as a MIMO radar that performs one-dimensional beam scanning in thehorizontal direction. However, in cases where the number of antennaelements is restricted, the aperture length is restricted in comparisonwith a MIMO radar performing one-dimensional scanning. Restricting theaperture length results in lower resolution in the horizontal directionas compared with a MIMO radar performing one-dimensional scanning.

Also, in order to realize MIMO radar with reduced probability oferroneous detection, the virtual receiving array preferably isconfigured so that there are less sidelobes in the beam being formed.

First Embodiment

According to an aspect of the present disclosure, MIMO radar can beconfigured with capable of three-dimensional measurement with additionalangular estimation capabilities in the vertical direction, whilesuppressing deterioration of angular separation capabilities in thehorizontal direction as compared with a MIMO radar performingone-dimensional beam scanning.

Embodiments of the present disclosure will be described below in detailwith reference to the drawings. Note that in the embodiments, componentsthat are the same are denoted by the same symbols, and redundantdescription thereof will be omitted.

Before describing the layout of multiple transmitting antennas(transmitting sub-arrays) and multiple receiving antennas (receivingsub-arrays), the configuration of the radar device will be described.Specifically, description will be made regarding the configuration of aMIMO radar where multiple transmitting antennas are switched by timedivision in the transmitting branch of the radar device, different radartransmission signals that have been time division multiplexed aretransmitted, and the transmission signals are separated and receptionprocessing is performed at the receiving branch. Note that theconfiguration of the radar device is not restricted to transmittingdifferent radar transmission signals that have been time divisionmultiplexed. For example, the radar transmission signals may befrequency division multiplexed instead of time division multiplexed orcode division multiplexed. That is to say, a configuration may be madewhere different radar transmission signals that have been frequencydivision multiplexed are transmitted by multiple transmitting antennasat the transmitting branch, and the transmission signals are separatedand reception processing is performed at the receiving branch.Similarly, a configuration of the radar device may be made where radartransmission signals that have been code division multiplexed aretransmitted by multiple transmitting antennas at the transmittingbranch, and reception processing is performed at the receiving branch.It should be noted that the embodiments described below are onlyexemplary, and that the present disclosure is not restricted by thefollowing embodiments.

Configuration of Radar Device 10

FIG. 1 is a block diagram illustrating an example of the configurationof a radar device 10 according to the present disclosure. The radardevice 10 has a radar transmitting unit (also referred to as atransmitting branch or radar transmission circuit) 100, a radarreceiving unit (also referred to as a receiving branch or radarreception circuit) 200, a reference signal generating unit (referencesignal generating circuit) 300, and a control unit (control circuit)400.

The radar transmitting unit 100 generates high-frequency (radiofrequency) radar signals (radar transmission signals) based on referencesignals received from the reference signal generating unit 300. Theradar transmitting unit 100 then transmits the radar transmissionsignals while switching multiple transmitting antenna elements #1through #Nt by time division.

The radar receiving unit 200 receives returning wave signals that areradar transmission signals reflected at a target (omitted fromillustration), using multiple receiving antenna elements #1 through #Na.The radar receiving unit 200 performs processing synchronously with theradar transmitting unit 100 by performing the processing operationsdescribed below, using reference signals received from the referencesignal generating unit 300. The radar receiving unit 200 performs signalprocessing on returning wave signals received at each receiving antennaelement #1 through #Na, and performs detection of at least whether ornot a target is present, or estimation of the direction thereof. Notethat a target is an object that is to be detected by the radar device10, and includes, for example, vehicles (including those with twowheels, three wheels, and four wheels) and people.

The reference signal generating unit 300 connected to each of the radartransmitting unit 100 and the radar receiving unit 200. The referencesignal generating unit 300 supplies reference signals to the radartransmitting unit 100 and radar receiving unit 200, to synchronize theprocessing at the radar transmitting unit 100 and radar receiving unit200.

The control unit 400 sets the pulse code generated by the radartransmitting unit 100, the phase to be set in variable beam control atthe radar transmitting unit 100 and the level of amplification ofsignals by the radar transmitting unit 100, for each radar transmissioncycle Tr. The control unit 400 outputs control signals instructing pulsecode (code control signals), control signals instructing phase (phasecontrol signals), and control signals instructing amplification level oftransmission signals (transmission control signals) to the radartransmitting unit 100. The control unit 400 also outputs outputswitching signals instructing timing of switching of transmittingsub-arrays #1 through #N of the radar transmitting unit 100 (switchingof output of radar transmission signals) to the radar receiving unit200.

Configuration of Radar Transmitting Unit 100

FIG. 2 is a block diagram illustrating an example of the configurationof the radar transmitting unit 100 according to the present disclosure.The radar transmitting unit 100 has a radar transmission signalgenerating unit (radar transmission signal generating circuit) 101, atransmission frequency conversion unit (transmission frequencyconversion circuit) 105, an electrical power distributor (electricalpower distribution circuit) 106, a transmission amplifying unit(transmission amplifying circuit) 107, and a transmission array antenna108.

Although the following description will be made illustrating theconfiguration of the radar transmitting unit 100 using a coded pulseradar, but this is not restrictive. For example, the radar transmittingunit 100 is similarly applicable to radar transmission signals usingfrequency-modulated continuous-wave (FM-CW) radar frequency modulation.

The radar transmission signal generating unit 101 generates a timingclock (clock signal) where a reference signal received from thereference signal generating unit 300 is multiplied by a predeterminednumber, and generates radar transmission signals based on the generatedtiming dock. The radar transmission signal generating unit 101repeatedly outputs radar transmission signals at the radar transmissioncycle Tr, based on code control signals from the control unit 400 ateach predetermined radar transmission cycle Tr.

Radar transmission signals are expressed byy(k _(t) ,M)=I(k _(T) ,M)+jQ(k _(t) ,M)where j represents an imaginary unit, k represents discrete time, M isan ordinal number of the radar transmission cycle, and I(k_(T), M) andQ(k_(T), M) respectively represent the in-phase component and quadraturecomponent of radar transmission signal (k_(T), M) at discrete time k_(T)in the M'th radar transmission cycle.

The radar transmission signal generating unit 101 includes a codegenerating unit (code generating circuit) 102, a modulating unit(modulation circuit) 103, and a low-pass filter (LPF) 104.

The code generating unit 102 generates code a_(n)(M) (where n=1, . . . ,L) of a code sequence having a code length of L (pulse code), in theM'th radar transmission cycle, based on code control signals at eachradar transmission cycle Tr. Pulse code that yields low-range sidelobeproperties is used for the code a_(n)(M) generated at the codegenerating unit 102. Examples of the code sequence include Barker code,maximum-length sequence code, and Gold code. The code a_(n)(M) generatedby the code generating unit 102 may all be of the same code, or may becode including different codes.

The modulating unit 103 subjects the code a_(n)(M) output from the codegenerating unit 102 to pulse modulation (amplitude modulation,amplitude-shift keying (ASK), or pulse shift keying) or phase modulation(phase-shift keying (PSK), and outputs the modulated code to the LPF104.

Of the modulated signals output from the modulating unit 103, signalcomponents of a predetermined restricted bandwidth and lower are outputto the transmission frequency conversion unit 105 as baseband radartransmission signals by the LPF 104.

The transmission frequency conversion unit 105 performs frequencyconversion of the baseband radar transmission signals output from theLPF 104 into radar transmission signals of a predetermined carrierfrequency (radio frequency (RF)) band.

The electrical power distributor 106 distributes theradio-frequency-band radar transmission signals output from thetransmission frequency conversion unit 105 over a count lilt, andoutputs to the transmission amplifying units 107.

The transmission amplifying units 107 (107-1 through 107-Nt) eitheramplify the radar transmission signals to be output to a predeterminedlevel and output, or turn transmission output off, based on transmissioncontrol signals for each radar transmission cycle Tr instructed from thecontrol unit 400.

The transmission array antenna 108 has alt transmitting antenna elements#1 through #Nt (108-1 through 108-Nt). The transmitting antenna elements#1 through #Nt are respectively connected to individual transmissionamplifying units 107-1 through 107-Nt, and transmit radar transmissionsignals output from the individual transmission amplifying units 107-1through 107-Nt.

FIG. 3 is a diagram illustrating an example of radar transmissionsignals according to the present disclosure. In each radar transmissioncycle Tr, the pulse code sequence is transmitted during a codetransmission slot Tw, and the remainder (Tr−Tw) is a non-transmissionslot. A pulse code sequence of code length L is included in the codetransmission slot Tw. An L count of sub-pulses is included in a singlecode. Also, pulse modulation is performed using an No count of samplesper sub-pulse, so there are Nr(No×L) samples in each code transmissionslot Tw. Further, an Nu count of samples is included in thenon-transmission slot (Tr−Tw) of the radar transmission cycle Tr.

FIG. 4 illustrates an example of time division switching operations ofthe transmitting antenna elements #1 through #Nt by the control unit 400according to the present disclosure. In FIG. 4, the control unit 400outputs control signals (code control signals, transmission controlsignals) to instruct switching the output from the transmission antennaelements, in order from the transmitting antenna element #1 through thetransmitting antenna element #Nt, to the radar transmitting unit 100 ateach radar transmission cycle Tr. With the transmission output durationof each transmitting sub-array as (Tr×Nb), the control unit 400 effectscontrol to repeat switching operations for the transmission outputduration for all transmitting sub-arrays, which is (Tr×Np)=(Tr×Nb×Nt),Nc times. The later-described radar receiving unit 200 performsmeasurement processing based on the switching operations of the controlunit 400.

For example, in a case of transmitting radar transmission signals fromthe transmitting antenna element #1, the control unit 400 outputs atransmission control signal to the transmission amplifying unit 107-1connected to the transmitting antenna element #1, instructingamplification of input signals to a predetermined level, and outputstransmission control signals to the transmission amplifying units 107-2through 107-Nt not connected to the transmitting antenna element #1,instructing to leave transmission output off.

In the same way, in a case of transmitting radar transmission signalsfrom the transmitting antenna element #2, the control unit 400 outputs atransmission control signal to the transmission amplifying unit 107-2connected to the transmitting antenna element #2, instructingamplification of input signals to a predetermined level, and outputstransmission control signals to the transmission amplifying units 107not connected to the transmitting antenna element #2, instructing toleave transmission output off. Thereafter the control unit 400 performsthe same control to the transmitting antenna elements #3 through #Nt inorder. This so far has been a description of output switching operationsof radar transmission signals by the control unit 400.

Another Configuration of Radar Transmitting Unit 100

FIG. 5 is a block diagram illustrating an example of anotherconfiguration of the radar transmission signal generating unit 101according to the present disclosure. The radar transmitting unit 100 mayhave, instead of the radar transmission signal generating unit 101, aradar transmission signal generating unit 101 a illustrated in FIG. 5.The radar transmission signal generating unit 101 a does not have thecode generating unit 102, modulating unit 103, and the LPF 104,illustrated in FIG. 2, and instated has a code storage unit (codestorage circuit) 111 and DA converting unit (DA converting circuit) 112illustrated in FIG. 5.

The code storage unit 111 stores a code sequence generated at the codegenerating unit 102 illustrated in FIG. 2 beforehand, and cyclicallyreads out the stored code sequence in order. The DA converting unit 112converts the code sequence (digital signals) output from the codestorage unit 111 into analog baseband signals.

Configuration of Radar Receiving Unit 200

FIG. 6 is a block diagram illustrating an example of the configurationof the radar receiving unit 200 according to the first and secondembodiments. The radar receiving unit 200 has a receiving array antenna202, an Na count of antenna element system processing units (antennaelement system processing circuits) 201 (201-1 through 201-Na), and adirection estimating unit (direction estimating circuit) 214.

The receiving array antenna 202 has an Na count of receiving antennaelements #1 through #Na (202-1 through 202-Na), The Na count ofreceiving antenna elements 202-1 through 202-Na receive returning wavesignals that are radar transmission signals that have reflected off of areflecting object including a measurement target (object), and outputsthe received returning wave signals to the respectively correspondingantenna element system processing units 201-1 through 201-Na asreception signals.

The antenna element system processing units 201 (201-1 through 201-Na)each have a receiving radio unit (receiving radio circuit) 203 and asignal processing unit (signal processing circuit) 207. The receivingradio unit 203 and signal processing unit 207 generate a timing clock(reference clock signal) where a reference signal received from thereference signal generating unit 300 is multiplied by a predeterminednumber, and operate based on the generated timing clock, therebyensuring synchronism with the radar transmitting unit 100.

The receiving radio units 203 each have an amplifying unit (amplifyingcircuit) 204, a frequency converter (frequency conversion circuit) 205,a quadrature detector (quadrature detecting circuit) 206. Specifically,at the z'th receiving radio unit 203, the amplifier 204 amplifiesreception signals received from the z'th receiving antenna element #z toa predetermined level, where z=1, . . . , Nr. Next, the frequencyconversion unit 205 performs frequency conversion of high-frequency-bandreception signals to baseband range. Thereafter, the quadrature detector206 converts the baseband range reception signals into baseband rangereception signals including I signals and Q signals.

The signal processing units 207 each have a first AD converting unit (ADconverting circuit) 208, a second AD converting unit (AD convertingcircuit) 209, a correlation computing unit (correlation computingcircuit) 210, an adding unit (adding circuit) 211, an output switchingunit (output switching circuit) 212, and an Nt count of doppleranalyzing units (doppler analyzing circuits) 213-1 through 213-Nt.

The first AD converting unit 208 inputs I signals from the quadraturedetector 206. The first AD converting unit 208 performs discrete-timesampling of baseband signals including I signals, thereby converting theI signals into digital data.

The second AD converting unit 209 inputs Q signals from the quadraturedetector 206. The second AD converting unit 209 performs discrete-timesampling of baseband signals including Q signals, thereby converting theQ signals into digital data.

Now, in the sampling by the first AD converting unit 208 and second ADconverting unit 209, an Ns count of discrete samples are taken per timeTp (i.e., Tw/L) of a single sub-pulse in radar transmission signals.That is to say, the oversampling count per sub-pulse is Ns.

FIG. 7 illustrates an example of transmission timing of radartransmission signals and measurement range of the radar device 10according to the present disclosure. In the following description, thebaseband reception signals at discrete time k in an M'th radartransmission cycle Tr[M] as the output of the first AD converting unit208 and second AD converting unit 209 are expressed as complex numbersignals x_(z)(k, M) M)=I_(z)(k, M)+jQ_(z)(k, M), using I signal I_(z)(k,M) and Q signal Q_(z)(k, M). Also note that in the following, thediscrete time k uses the timing as which the radar transmission cycle(Tr) starts as a reference (k=1), and the signal processing unit 207cyclically performs measurement up to a sample point k (N_(r)+N_(u))N_(s)/N_(o) before the radar transmission cycle Tr ends. That is to say,k=1, . . . , (N_(r)+N_(u)) N_(s)/N_(o) holds. Also, j is an imaginaryunit here.

At the z'th signal processing unit 207, the correlation computing unit210 performs correlation computation between discrete sample valuex_(z)(k, M) received from the first AD converting unit 208 and second ADconverting unit 209, and pulse code a_(n)(M) of a code length L (wherez=1, . . . , Na, and n=1, . . . , L) transmitted by the radartransmitting unit 100, for each radar transmission cycle Tr. Forexample, the correlation computing unit 210 performs sliding correlationcomputation between discrete sample value x_(z)(k, M) and pulse codea_(n)(M). For example, correlation computation value AC_(z)(k, M) fromsliding correlation computation of discrete time k at the M'th radartransmission cycle Tr[M] is calculated based on Expression (1)AC _(z)(k,M)=Σ_(n=1) ^(L) x _(z)(k+N _(s)(n−1),M)a _(n)(M)*  (1)where the asterisk “*” represents a complex conjugate operator.

The correlation computing unit 210 performs correlation computation overa period k=1, . . . , (N_(r)+N_(u)) N_(s)/N_(o), for example, inaccordance with Expression (1). Note however, that the correlationcomputing unit 210 is not restricted to cases of performing correlationcomputation over a period k=1, . . . , (N_(r)+N_(u)) N_(s)/N_(o), andthat the measurement range (I.e., the range of k) may be restricted inaccordance with the range where the target that is the object ofmeasurement by the radar device 10 is present. This restriction reducesthe amount of computation processing at the correlation computing unit210. For example, the measurement range of the correlation computingunit 210 may be restricted to k=N_(s) (L+1), . . . , (N_(r)+N_(u))N_(s)/N_(o)−N_(s)L. In this case, the radar device 10 does not performmeasurement in a time slot corresponding to code transmission slot Tw,as illustrated in FIG. 7.

According to the above-described configuration, even in a case wherethere is direct leakage of radar transmission signals at the radarreceiving unit 200, processing by the correlation computing unit 210 isnot performed during the period where there is leakage of radartransmission signals (at least during a period below τ1). Accordingly,the radar device 10 is capable of measurement with the effects ofleakage eliminated. Also, in a case of restricting the measurement range(range of k), processing can be applied in the same way with themeasurement range (range of k) restricted with regard to the processingat the adding unit 211, output switching unit 212, doppler analyzingunit 213, and direction estimating unit 214, which will be describedbelow. Accordingly, the amount of processing at each component can berestricted, and the power consumption of the radar receiving unit 200can be reduced.

At the z'th signal processing unit 207, the adding unit 211 performsaddition (coherent integration) processing using the correlationcomputation value AC_(z)(k, M) received from the correlation computingunit 210 each discrete time k, with a period (Tr×Nb) of multiple timesNb in radar transmission cycles Tr continuously transmitted from theN_(D)'th transmitting antenna element #N_(D) as an increment, based onoutput switching signals output from the control unit 400. Note thatN_(D)=1, . . . , Nt, and z=1, . . . , Na here.

The addition processing (coherent integration) spanning the period(Tr×Nb) is represented by the following Expression (2)CI_(z) ^((N) ^(D) ⁾(k,m)=Σ_(g=1) ^(N) ^(b) AC _(z)(k,(N×N _(b))(m−1)+(N_(D)−1)×N _(b) +g)  (2)where CI_(z) ^((ND))(k, m) represents the addition value of correlationcomputation values (hereinafter referred to as correlation additionvalue), m is an integer of 1 or greater indicating an ordinal number ofthe number of times of addition at the adding unit 211, and z=1, . . . ,Na.

In order to obtain ideal addition gain is for the phase component ofcorrelation computation values to be preferably aligned to a certainextent in the addition range of correlation computation values. That isto say, the number of times of addition preferably is set based on theestimated greatest velocity of movement of the target that is the objectof measurement. The reason is that the greater the velocity of movementof the target is, the greater the variation of doppler frequencycontained in waves reflected from the target is, and the temporal periodwhere there is a high correlation is short, so Np (i.e., N×Nb) is asmall value, and the advantages of increased gain by addition at theadding unit 211 decreases.

At the z'th signal processing unit 207, the output switching unit 212selectively switches and outputs CI_(z) ^((ND))(k, m) to the N_(D)'thdoppler analyzing unit 213-N_(D), based on output switching signalsoutput from the control unit 400. Note that CI_(z) ^((ND))(k, m) is theaddition results for each discrete time k, where addition has beenperformed in increments of multiple Nb periods (Tr×Nb) of radartransmission cycles Tr continuously transmitted from the N_(D)'thtransmitting antenna element. Note that N_(D)=1, . . . , Nt, and z=1 . .. , Na, here.

The signal processing unit 207 has doppler analyzing unit 213-1 through213-Nt, of the same count Nt as the transmitting antenna elements #1through #Nt. The doppler analyzing unit 213 (213-1 through 213-Nt)performs coherent integration with the timing of discrete time kaligned, with CI_(z) ^((ND))(k, N_(C)(w−1)+1) through CI_(z) ^((ND))(k,N_(C)×w) that is the output of a count N_(C) of adding units 211obtained each discrete time k as an increment. For example, the doppleranalyzing unit 213 performs coherent integration after having correctedphase variation Φ(f_(s))=2πf_(s)(T_(r)×N_(b)) ΔΦ in accordance with acount 2Nf of different doppler frequencies f_(s)ΔΦ as shown in thefollowing Expression (3)

$\begin{matrix}{{{FT\_ CI}_{z}^{(N_{D})}\left( {k,f_{s},w} \right)} = {{\sum\limits_{q = 0}^{N_{c} - 1}\;{{{CI}_{z}^{(N_{D})}\left( {k,{{N_{c}\left( {w - 1} \right)} + q + 1}} \right)}{\exp\left\lbrack {{- j}\;{\Phi\left( f_{s} \right)}q} \right\rbrack}}} = {\sum\limits_{q = 0}^{N_{c} - 1}\;{{{CI}_{z}^{(N_{D})}\left( {k,{{N_{c}\left( {w - 1} \right)} + q + 1}} \right)}{\exp\left\lbrack {{- j}\; 2\pi\; f_{s}T_{r}N_{b}q\;{\Delta\Phi}} \right\rbrack}}}}} & (3)\end{matrix}$where FT_CI_(z) ^((ND))(k, f_(s), w) is the w'th output at the N_(D)'thdoppler analyzing unit 213-N_(D) in the z'th signal processing unit 207,and indicates the results of coherent integration of doppler frequenciesf_(s)ΔΦ at discrete time k, with regard to the N_(D)'th output of theadding unit 211. Note that N_(D)=1, . . . , Nt holds, f_(s)=−Nf+1, . . ., 0, Nf holds, k=1, . . . , (Nr+Nu) Ns/No holds, w is a natural number,ΔΦ is a phase rotation unit, j is an imaginary unit, and z=1, . . . , Naholds.

Accordingly, the signal processing unit 207 can obtain FT_CI_(z)^((ND))(k, −Nf+1, w), . . . , FT_CI_(z) ^((ND))(k, Nf−1, w) that is theresults of coherent integration in accordance with a count 2Nf ofdoppler frequency components every discrete time K each period(Tr×Nb×Nc) of multiple times Nb×Nc of radar transmission cycles Tr.

In a case where ΔΦ=1/N_(c), the processing at the doppler analyzing unit213 described above is equivalent to performing discrete Fouriertransform (DFT) processing of the output of the adding unit 211 at asampling interval T_(m)=(Tr×N_(p)) and a sampling frequencyf_(m)=1/T_(m).

Setting Nf to be a number that is a power of 2 enables the doppleranalyzing unit 213 to apply fast Fourier transform (FFT) processing, andthe amount of computation processing can be reduced. Note that whenNf>Nc, performing zero padding processing where CI_(z) ^((ND))(k,Nc(w−1)+1)=0 in the region where q>Nc holds similarly enables FFTprocessing to be applied, and the amount of computation processing canbe reduced.

Also, the doppler analyzing unit 213 may perform processing ofsuccessively computing the multiply-add operation in Expression (3)above, instated of FFT processing. That is to say, the doppler analyzingunit 213 may generate a coefficient exp[−j2πf_(s)T_(r)N_(b)qΔΦ]corresponding to f_(s)=−Nf+1, . . . , 0, Nf, with regard to CI_(z)^((ND))(k, Nc(w−1)+q+1) that is Nc outputs of the adding unit 211obtained at each discrete time k, and successively perform multiply-addprocessing. Note that q=0 . . . . , N_(c)−1 here.

Note that in the following description, the w'th output FT_CI_(z) ⁽¹⁾(k,f_(s), w), . . . , FT_CI_(z) ^((Na))(k, f_(s), w), obtained byperforming the same processing on each of the signal processing unit 207of the first antenna element system processing unit 201-1 through theNa'th antenna element system processing unit 201-Na, will be written asa virtual receiving array correlation vector h(k, f_(s), w) in thefollowing Expression (4) (or Expression (5)).

$\begin{matrix}{{h\left( {k,{fs},w} \right)} = {\quad{\left\lbrack \begin{matrix}{{FT\_ CI}_{1}^{(1)}\left( {k,f_{s},w} \right)} \\{{FT\_ CI}_{1}^{(2)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\; T_{r}N_{b}} \right)}} \\\vdots \\{{FT\_ CI}_{1}^{(N)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\;\left( {N - 1} \right)T_{r}N_{b}} \right)}} \\{{FT\_ CI}_{2}^{(1)}\left( {k,f_{s},w} \right)} \\{{FT\_ CI}_{2}^{(2)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\; T_{r}N_{b}} \right)}} \\\vdots \\{{FT\_ CI}_{2}^{(N)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\;\left( {N - 1} \right)T_{r}N_{b}} \right)}} \\\vdots \\{{FT\_ CI}_{Na}^{(1)}\left( {k,f_{s},w} \right)} \\{{FT\_ CI}_{Na}^{(2)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\; T_{r}N_{b}} \right)}} \\\vdots \\{{FT\_ CI}_{Na}^{(N)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\;\left( {N - 1} \right)T_{r}N_{b}} \right)}}\end{matrix} \right\rbrack = \left\lbrack \begin{matrix}{h_{1}\left( {k,{fs},w} \right)} \\{h_{2}\left( {k,{fs},w} \right)} \\\vdots \\{h_{Na}\left( {k,{fs},w} \right)}\end{matrix} \right\rbrack}}} & (4) \\{{h_{Z}\left( {k,{fs},w} \right)} = {\quad\left\lbrack \begin{matrix}{{FT\_ CI}_{z}^{(1)}\left( {k,f_{s},w} \right)} \\{{FT\_ CI}_{z}^{(2)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\; T_{r}N_{b}} \right)}} \\\vdots \\{{FT\_ CI}_{z}^{(N)}\left( {k,f_{s},w} \right){\exp\left( {{- j}\; 2\pi\; f_{s}\;{\Delta\Phi}\;\left( {N - 1} \right)T_{r}N_{b}} \right)}}\end{matrix} \right\rbrack}} & (5)\end{matrix}$

The virtual receiving array correlation vector h(k, f_(s), w) includesNt×Na elements, being the product of the count Nt of transmittingantenna elements #1 through #Nt and the count Na of receiving antennaelements #1 through #Na. This virtual receiving array correlation vectorh(k, f_(s), w) is used in description of processing where directionestimation processing is performed based on phase difference among thereceiving antenna elements #1 through #Na as to returning wave signalsfrom the target, which will be described later. Note that z=1, . . . ,Na and N_(D)=1, . . . , Nt here.

In the above-described Expressions (4) and (5), phase rotation for eachdoppler frequency (f_(s)ΔΦ), due to difference in transmission time fromthe transmitting sub-arrays, is corrected. That is to say, receptionsignal FT_CI_(z) ^((Na))(k, f_(s), w) of the doppler frequency (f_(s)ΔΦ)component from the N_(D)'th transmitting sub-array is multiplied by exp[−j2πf_(s)ΔΦ(N_(D)−1)T_(f)N_(b)], with the first transmitting sub-array(N_(D)=1) as a reference. This so far has been description of processingby the components of the signal processing unit 207.

The direction estimating unit 214 calculates a virtual receiving arraycorrelation vector h_(_after_cal)(k, f_(s), w) where inter-antennadeviation has been corrected, by multiplying the virtual receiving arraycorrelation vector h (k, f_(s), w) of the w'th doppler analyzing unit213 output from the signal processing unit 207 of the first antennaelement system processing unit 201-1 through the signal processing unit207 of the Na'th antenna element system processing unit 201-Na, by anarray correction value h_(cal[b]) that corrects phase-shift deviationand amplitude deviation between the transmission array antenna 108 andthe receiving array antenna 202, as in the following Expression (6).Note that b=1, . . . , (Nt×Na) here.

$\begin{matrix}{{{h_{{\_{after}}{\_{cal}}}\left( {k,{fs},w} \right)} = {{{CA}\mspace{14mu}{h\left( {k,{fs},w} \right)}} = \left\lbrack \begin{matrix}{h_{1}\left( {k,{fs},w} \right)} \\{h_{2}\left( {k,{fs},w} \right)} \\\vdots \\{h_{{Na} \times {Nr}}\left( {k,{fs},w} \right)}\end{matrix} \right\rbrack}}{{CA} = \begin{bmatrix}{h\_ cal}_{\lbrack 1\rbrack} & 0 & \ldots & 0 \\0 & {h\_ cal}_{\lbrack 2\rbrack} & \ldots & 0 \\\vdots & \ddots & \ddots & \vdots \\0 & \ldots & 0 & {h\_ cal}_{\lbrack{{Nt} \times {Na}}\rbrack}\end{bmatrix}}} & (6)\end{matrix}$

The virtual receiving array correlation vector h_(_after_cal)(k, f_(s),w) where inter-antenna deviation has been corrected is a column vectormade up of Na×Nr elements. The elements of the virtual receiving arraycorrelation vector h_(_after_cal)(k, f_(s), w) will be written as h₁(k,f_(s), w), . . . , h_(Na×Nr)(k, fs, w) and used in description ofdirection estimating processing below.

The direction estimating unit 214 then uses the virtual receiving arraycorrelation vector h_(_after_cal)(k, f_(s), w) to perform estimationprocessing of the direction of arrival of reflected wave signals, basedon the phase difference of reflected wave signals among the receivingantenna elements 202-1 through 202-Na. The direction estimating unit 214calculates a spatial profile with an azimuth θ in a direction estimationevaluation function value P_(H)(θ, k, fs, w) variable over apredetermined angular range, extracts a predetermined number of maximalpeaks in the calculated spatial profile, in order from the largest, andthe directions of the maximal peaks are taken as estimation values ofthe direction of arrival.

There are various types of the estimation evaluation function valueP_(H)(θ, k, fs, w) depending on the direction of arrival estimationalgorithm. For example, an estimation method using an array antenna thatis disclosed in “Direction-of-arrival estimation using signal subspacemodeling”, Cadzow. J. A., Aerospace and Electronic Systems, IEEETransactions on Volume: 28, Issue: 1 Publication Year: 1992, Page(s):64-79, may be used.

Beamforming, for example, can be expressed as in the followingExpressions (7) and (8)

$\begin{matrix}{{P_{H}\left( {\theta_{u},k,{fs},w} \right)} = {{{a_{H}\left( \theta_{u} \right)}^{H}{h_{{\_{after}}{\_{cal}}}\left( {k,{fs},w} \right)}}}^{2}} & (7) \\{{a_{H}\left( \theta_{u} \right)} = \begin{bmatrix}1 \\{\exp\left\{ {{- j}\; 2\pi\; d_{H}\sin\;{\theta_{u}/\lambda}} \right\}} \\\vdots \\{\exp\left\{ {{- j}\; 2{\pi\left( {N_{VAH} - 1} \right)}\; d_{H}\sin\;{\theta_{u}/\lambda}} \right\}}\end{bmatrix}} & (8)\end{matrix}$where the superscript H is a Hermitian transpose operator, a_(H)(θ_(u))indicates the direction vector of the virtual receiving array as towaves arriving from azimuth θ_(u), and θ_(u) is changed by apredetermined direction interval β₁ within the range of direction inwhich arrival direction estimation is performed. This θ_(u) is, forexample, set as followsθ_(u)=θ min+uβ ₁ ,u=0, . . . ,NUNU=floor[(θ max−θ min)/β₁]+1where floor(x) is a function that returns the largest integer value thatdoes not exceed real number x. Note that techniques such as Capon,MUSIC, and so forth may be applied in the same way, instead ofbeamforming.

FIG. 8 illustrates a three-dimensional coordinates system used fordescription of operations of the direction estimating unit 214 accordingto the present disclosure. Description will be made below regarding acase where estimation processing is made in two-dimensional directions,by applying the processing of the direction estimating unit 214 to thethree-dimensional coordinates system illustrated in FIG. 8.

In FIG. 8, the positional vector of target P_(T) is defined as r_(PT),with the origin O as a reference. Also, a projection point where thepositional vector r_(PT) of the target P_(T) is projected on the XZplane is P_(T)′ in FIG. 8. In this case, the azimuth θ is defined as anangle formed between line O-P_(T)′ and the Z axis (θ>0 in a case wherethe X coordinate of target P_(T) is positive). Elevation Φ is defined asthe angle between the target P_(T) and a line connecting the origin Oand projected point P_(T)′, within a plane including the target P_(T),the origin O, and projected point P_(T)′ (Φ>0 in a case where the Ycoordinate of the target P_(T) is positive). Note that description willbe made below with a case where the transmission array antenna 108 andreceiving array antenna 202 are disposed within the XY plane, as oneexample.

The positional vector of the n_(va)'th antenna element in the virtualreceiving array with the origin O as a reference is written as Sn_(va).Note that n_(va)=1, . . . , Nt×Na here.

The positional vector S₁ of the first (n_(va)=1) antenna element in thevirtual receiving array is decided based on the positional relationbetween physical position of the first receiving antenna element Rx #1and the origin O. The positional vectors S₂, . . . , Sn_(va), of theother antenna elements in the virtual receiving array are decided in astate where the relative layout of the virtual receiving array, decidedfrom the inter-element spacing of the transmission array antenna 108 andreceiving array antenna 202 in the XY plane, is maintained, with thepositional vector S₁ of the first antenna element as a reference. Notethat the origin O may be made to coincide with the physical position ofthe first receiving antenna element Rx #1.

In a case of the radar receiving unit 200 receiving reflected waves froma target P_(T) at a far field, the phase difference d(r_(PT), 2, 1) ofreception signals at the second antenna element, with reception signalsat the first antenna element of the virtual receiving array as areference, is as shown in the following Expression (9)

$\begin{matrix}{{d\left( {r_{PT},2,1} \right)} = {{{- \frac{2\;\pi}{\lambda}}\frac{\left\langle {{- r_{PT}},\left( {S_{2} - S_{1}} \right)} \right\rangle}{r_{PT}}} = {{\frac{2\;\pi}{\lambda}\left\langle {\frac{r_{PT}}{r_{PT}},\left( {S_{2} - S_{1}} \right)} \right\rangle} = {\frac{2\;\pi}{\lambda}\left\langle {\frac{r_{PT}}{r_{PT}},{D\left( {2,1} \right)}} \right\rangle}}}} & (9)\end{matrix}$where <x, y> is an inner product operator of vector x and vector y.

The positional vector of the second antenna element, with the positionalvector of the first antenna element of the virtual receiving array as areference, is shown in the following expression (10) as inter-elementvector D(2, 1).D(2,1)=S ₂ −S ₁  (10)

In the same way, in a case of the radar receiving unit 200 receivingreflected waves from a target P_(T) at a far field, the phase differenced(r_(PT), n_(va) ^((t)), n_(va) ^((r))) of reception signals at then_(va) ^((t))'th antenna element, with reception signals at the n_(va)^((r))'th antenna element of the virtual receiving array as a reference,is as shown in the following Expression (11)

$\begin{matrix}{{d\left( {r_{PT},n_{va}^{(t)},n_{va}^{(r)}} \right)} = {\frac{2\;\pi}{\lambda}\left\langle {\frac{r_{PT}}{r_{PT}},{D\left( {n_{va}^{(t)},n_{va}^{(r)}} \right)}} \right\rangle}} & (11)\end{matrix}$where n_(va) ^((r))=1, . . . , Nt×Na holds and n_(va) ^((t))=1, . . . ,Nt×Na holds.

The positional vector of the n_(va) ^((t))'th antenna element, with thepositional vector of the n_(va) ^((r))'th antenna element of the virtualreceiving array as a reference, is shown in the following expression(12) as inter-element vector D(n_(va) ^((t)), n_(va) ^((r))).D(n _(va) ^((t)) ,n _(va) ^((r)))=S _(n) _(va) _((t)) −S _(n) _(va)_((r))   (12)

The phase difference d(r_(PT), n_(va) ^((t)), n_(va) ^((r))), n_(va)^((r))) of reception signals at the n_(va) ^((t))'th antenna element,with reception signals at the n_(va) ^((r))'th antenna element of thevirtual receiving array as a reference, is dependent on unit vector(r_(PT)/|r_(PT)|) indicating the direction of the target P_(T) at a farfield and the inter-element vector D(n_(va) ^((t)), n_(va) ^((r))), asshown in the above-described Expressions (11) and (12).

Also, in a case where the virtual receiving array exists within the sameplane, the inter-element vector D(n_(va) ^((t)), n_(va) ^((r))) existson the same plane. The direction estimating unit 214 uses part or all ofsuch inter-element vectors to configure a virtual plane layout arrayantenna assuming that antenna elements virtually exist at positionsindicated by the inter-element vectors, and performs two-dimensionaldirection estimation processing. That is to say, the directionestimating unit 214 performs direction of arrival estimation processingusing multiple virtual antennas interpolated by interpolation processingwith regard to antenna elements making up the virtual receiving array.

In a case where virtual antenna elements overlap, the directionestimating unit 214 may fixedly select one of the overlapping antennaelements beforehand. Alternatively, the direction estimating unit 214may perform averaging processing using reception signals at alloverlapping virtual antenna elements.

Description will be made below regarding two-dimensional directionestimation processing using beamforming, in a case where the virtualplane layout array antenna has been configured using a N_(q) count ofinter-element vector groups. The nq'th inter-element vector making upthe virtual plane layout array antenna will be written as D(n_(va(nq))^((t)), n_(va(nq)) ^((r))), where nq=1, . . . , Nq holds.

The direction estimating unit 214 generates the virtual plane layoutarray antenna correlation vector h_(VA)(k, fs, w) shown in Expression(13) below, using h₁(k, fs, w), . . . , h_(Na×N)(k, fs, w) that areelements of the virtual receiving array correlation vectorh_(_after_cal)(k, fs, w).

$\begin{matrix}{{h_{VA}\left( {k,{fs},w} \right)} = {{{CA}\mspace{14mu}{h\left( {k,{fs},w} \right)}} = {\quad\left\lbrack \begin{matrix}{{h_{n_{{va}{(1)}}^{(t)}}\left( {k,{fs},w} \right)}{{h_{n_{{va}{(1)}}^{(r)}}^{*}\left( {k,{fs},w} \right)}/{{h_{n_{{va}{(1)}}^{(r)}}^{*}\left( {k,{fs},w} \right)}}}} \\{{h_{n_{{va}{(2)}}^{(t)}}\left( {k,{fs},w} \right)}{{h_{n_{{va}{(2)}}^{(r)}}^{*}\left( {k,{fs},w} \right)}/{{h_{n_{{va}{(2)}}^{(r)}}^{*}\left( {k,{fs},w} \right)}}}} \\\vdots \\{{h_{n_{{va}{(N_{q})}}^{(t)}}\left( {k,{fs},w} \right)}{{h_{n_{{va}{(N_{q})}}^{(r)}}^{*}\left( {k,{fs},w} \right)}/{{h_{n_{{va}{(N_{q})}}^{(r)}}^{*}\left( {k,{fs},w} \right)}}}}\end{matrix} \right\rbrack}}} & (13)\end{matrix}$

The following Expression (14) shows a virtual plane layout arraydirection vector a_(VA)(θu, Φv).

$\begin{matrix}{{a_{VA}\left( {\theta_{u},\Phi_{v}} \right)} = \begin{bmatrix}{\exp\left\{ {j\frac{2\pi}{\lambda}\left\langle {\frac{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}{{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}},{D\left( {n_{{va}{(1)}}^{(t)},n_{{va}{(1)}}^{(r)}} \right)}} \right\rangle} \right\}} \\{\exp\left\{ {j\frac{2\pi}{\lambda}\left\langle {\frac{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}{{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}},{D\left( {n_{{va}{(2)}}^{(t)},n_{{va}{(2)}}^{(r)}} \right)}} \right\rangle} \right\}} \\\vdots \\{\exp\left\{ {j\frac{2\pi}{\lambda}\left\langle {\frac{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}{{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}},{D\left( {n_{{va}{(N_{q})}}^{(t)},n_{{va}{(N_{q})}}^{(r)}} \right)}} \right\rangle} \right\}}\end{bmatrix}} & (14)\end{matrix}$

The relation between the unit vector (r_(PT)/|r_(PT)|) indicating thedirection of the target P_(T), and the azimuth θ and the elevation Φ, ina case where the virtual receiving array is in the XY plane, is as shownin the following Expression (15).

$\begin{matrix}{\frac{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}{{r_{PT}\left( {\theta_{u},\Phi_{v}} \right)}} = \begin{pmatrix}{\sin\mspace{14mu}\theta_{u}\mspace{14mu}\cos\;\Phi_{v}} \\{\sin\mspace{14mu}\Phi_{v}} \\{\cos\mspace{14mu}\theta_{u}\mspace{14mu}\cos\;\Phi_{v}}\end{pmatrix}} & (15)\end{matrix}$

The direction estimating unit 214 calculates the unit vector(r_(PT)/|r_(PT)|) using the above Expression (15), for azimuth θu,elevation Φv regarding which the two-dimensional spatial profile in thevertical direction and horizontal direction is to be calculated.Further, the direction estimating unit 214 performs two-dimensionaldirection estimation processing in the horizontal direction and verticaldirection, using the virtual plane layout array antenna correlationvector h_(VA)(k, fs, w) and virtual plane layout array direction vectora_(VA)(θu, Φv).

For example, in two-dimensional direction estimation processing usingbeamforming, the two-dimensional spatial profile in the verticaldirection and horizontal direction is calculated using an evaluationfunction for two-dimensional direction estimation that is shown in thefollowing Expression (16) using the virtual plane layout array antennacorrelation vector h_(VA)(k, fs, w) and virtual plane layout arraydirection vector a_(VA)(θu, Φv). The azimuth and elevation exhibitingthe greatest value or maximal value in the two-dimensional spatialprofile is taken to be the estimation value for the direction ofarrival.P _(BA)(θ_(u),Φ_(v) ,k,fs,w)=|a _(VA)(θ_(u),Φ_(v))^(N) h_(VA)(k,fs,w)|²  (16)

Note that the direction estimating unit 214 may apply a high-resolutiondirection of arrival estimation algorithm such as Capon, MUSIC, or thelike, using the virtual plane layout array antenna correlation vectorh_(VA)(k, fs, w) and virtual plane layout array direction vectora_(VA)(θu, Φv), instead of beamforming. This increases the computationamount, but angular resolution can be raised.

Also, the above-described discrete time k may be output converted intodistance information. The following Expression (17) can be used toconvert the discrete time k into distance information R(k)

$\begin{matrix}{{R(k)} = {k\frac{T_{w}C_{0}}{2\; L}}} & (17)\end{matrix}$where Tw represents code transmission slot, L represents pulse codelength, and C₀ represents the speed of light.

Further, doppler frequency information may be converted into relativespeed component and output. The following Expression (18) can be used toconvert the doppler frequency fsΔΦ into relative speed componentv_(d)(f_(s))

$\begin{matrix}{{v_{d}\left( f_{s} \right)} = {\frac{\lambda}{2}f_{s}{\Delta\Phi}}} & (18)\end{matrix}$where λ is the wavelength of the carrier frequency of RF signals outputfrom the transmission frequency conversion unit 105.Antenna Element Layout in Radar Device 10 According to First Embodiment

The layout of the lit transmitting antenna elements Tx #1 through #Nt ofthe transmission array antenna 108 and the Na receiving antenna elementsRx #1 through #Na of the receiving array antenna 202, in the radardevice 10 having the above configuration, will be described below.

First Layout Example

FIG. 9 illustrates a first layout example of receiving antenna elementsRx #1 through Rx #Na of the receiving array antenna 202 according to thefirst embodiment. As illustrated in FIG. 9, the Na receiving antennaelements Rx #1 through Rx #Na of the receiving array antenna 202 arelaid out following the first axis. The Na receiving antenna elements Rx#1 through Rx #Na are basically laid out equidistantly at a firstspacing d_(H), with a part thereof being laid out at a third spacing duthat differs from the first spacing d_(H). In other words, of the #Na−1spacings that are the spacings among the Na adjacent receiving antennaelements Rx #1 through Rx #Na, part are equal to the third spacing duthat differs from the first spacing, and the remainder are equal to thefirst spacing d_(H).

FIG. 10A illustrates the first layout example of receiving antennaelements Rx #1 through Rx #8 of the receiving array antenna 202according to the first embodiment. In the example illustrated in FIG.10A, the receiving array antenna 202 has eight receiving antennaelements Rx #1 through Rx #8. Out of the eight receiving antennaelements Rx #1 through Rx #8, the receiving antenna elements Rx #1through Rx #7 are disposed equidistantly at the first spacing d_(H)following the first axis. The first spacing d_(H) here is equal to 0.5wavelengths, for example. The remaining receiving antenna element Rx #8is disposed at the third spacing 2×d_(H) from the receiving antennaelement Rx #7. That is to say, part of the receiving antenna elements Rx#1 through Rx #8 is laid out non-equidistantly. A synthesized aperturelength dRx at the phase centers of the receiving antenna elements Rx #1through Rx #8 is equal to 8×d_(H), which is the width of the receivingantenna elements Rx #1 through Rx #8 along the first axis.

FIG. 10B illustrates the first layout example of transmitting antennaelements Tx #1 and Tx #2 of the transmission array antenna 108 accordingto the first embodiment. In the example illustrated in FIG. 10B, thetransmission array antenna 108 has two transmitting antenna elements Tx#1 and Tx #2. The transmitting antenna elements Tx #1 and Tx #2 are laidout at a spacing of 8×d_(H) in a first axis direction, and a spacing ofd_(V) in a second axis direction that is orthogonal to the first axisdirection.

In one example, the first spacing d_(H) and the second spacing of d_(V)each may be 0.3 wavelengths or longer but two wavelengths or shorter,and may be around one-half wavelength. For example, the first spacingd_(H) and the second spacing of d_(V) may be equal to 0.5 wavelengths,with the wavelength used for radar transmission signals as a reference.

The first axis and second axis may be on the XY plane illustrated inFIG. 8, or may be disposed orthogonal to each other. For example, thefirst axis direction is the horizontal direction, and the second axisdirection is the vertical direction. Description will be made below withthe first axis direction agreeing with the horizontal direction and thesecond axis direction agreeing with the vertical direction, for the sakeof ease of description.

In a case of using the transmission array antenna 108 illustrated inFIG. 10B in a usage of long-range measurement to measure ahead of avehicle on a freeway, for example, the field of view (FoV) may benarrowed down to a narrow angle in the horizontal direction and verticaldirection. The FoV may be around 30 degrees in the horizontal directionand around 10 degrees in the vertical direction, for example.

The aperture length of each receiving antenna element of the receivingarray antenna 202 can be widened in the second axis direction and thebeam width in the vertical direction be narrowed down, with the pointsillustrated in FIG. 10A (hatched circles) as the phase centers, therebyobtaining high antenna gain. Also, the aperture length of eachtransmitting antenna element of the transmission array antenna 108 canbe widened in the first axis direction and second axis direction and thebeam width in the horizontal direction and the vertical direction benarrowed down, with the points illustrated in FIG. 10B (blank circles)as the phase centers, thereby obtaining high antenna gain. Note that theantenna elements may be configured using sub-array antennas, and anarray weight may be applied to the sub-array antenna to suppresssidelobes.

Note that dummy antenna elements may be disposed for the receivingantenna elements Rx #1 through Rx #8 laid out non-equidistantly asillustrated in FIG. 10A. A dummy antenna element is an antennaconfigured of antenna elements that have the same physical configurationas the other antenna elements, but is not used for transmission orreception of radar signals. For example, dummy antenna elements may bedisposed in non-equidistant regions such as between the receivingantenna element Rx #7 and the receiving antenna element Rx #8, a regionto the left side of the receiving antenna element Rx #1, or a region tothe right side of the receiving antenna element Rx #8. Disposing dummyantenna elements yields the advantages of uniform effects of electricproperties such as antenna radiation, impedance matching, and isolation,for example.

FIG. 100 illustrates a layout of a virtual receiving array according tothe first layout example. It can be seen from FIG. 100 that a pair ofvirtual antenna elements VA #8 and VA #9 are laid out adjacently at aspacing of d_(V) in the second axis direction. It can also be seen fromFIG. 100 that the aperture length in the first axis direction of thevirtual receiving array is equal to 16×d_(H).

A two-dimensional beam is configured by a two-dimensional virtualreceiving array extending in the first axis direction and the secondaxis direction. The fact that the layout of the receiving antennaelements Rx #1 through Rx #8 and transmitting antenna elements Tx #1 andTx #2 according to the first embodiment is an antenna layout that hashigh resolution in the horizontal direction and has angular estimationcapabilities in the vertical direction will be demonstrated below withby way of a first comparative example and second comparative example.

First Comparative Example

FIG. 11A illustrates the layout of the receiving antenna elements Rx #1through Rx #8 of the receiving array antenna according to the firstcomparative example. For the sake of comparison, the number of elementsin the receiving antenna elements Rx #1 through Rx #8 illustrated inFIG. 11A is equal to the number of elements in the receiving antennaelements Rx #1 through Rx #8 illustrated in FIG. 10A. The receivingantenna elements Rx #1 through Rx #8 are laid out equidistantly at thefirst spacing d_(H) in the first axis direction, as illustrated in FIG.11A.

FIG. 11B illustrates the layout of the transmitting antenna elements Tx#1 and Tx #2 of the transmitting array antenna according to the firstcomparative example. For the sake of comparison, the number of elementsin the transmitting antenna elements Tx #1 and Tx #2 illustrated in FIG.11B is equal to the number of elements in the transmitting antennaelements Tx #1 and Tx #2 illustrated in FIG. 10B. The transmittingantenna elements Tx #1 and Tx #2 are laid out at a spacing of 7×d_(H) inthe first axis direction, and at a spacing of d_(V) in the second axisdirection that is orthogonal to the first axis direction so that atleast one part of the virtual receiving arrays is arrayed in the secondaxis direction.

FIG. 11C illustrates the layout of a virtual receiving array accordingto the first comparative example. It can be seen in FIG. 11C that in thevirtual receiving array, VA #8 and VA #9 are laid out adjacently at aspacing of d_(V) in the second axis direction, in the same way asillustrated in FIG. 10C. However, the aperture length in the first axisdirection of the virtual receiving array is 14×d_(H), which is smallerthan the aperture length in the first axis direction of the virtualreceiving array according to the first embodiment that is 16×d_(H).

FIG. 12 is a cross sectional view of two-dimensional beams according tothe first layout example and the first comparative example at 0 degreesin the second axis direction, taken along the first axis direction. Thebeam width 6.2 degrees corresponding to −3 dB according to the firstembodiment is smaller than the beam width of 7.0 degrees according tothe first comparative example as illustrated in FIG. 12. That is to say,the antenna layout according to the first embodiment yields higherresolution in the horizontal direction than the antenna layout accordingto the first comparative example. Note that when the field of view isnarrowed in the horizontal direction, the magnitude of sidelobes at thewide-angle side do not substantially affect the probability of erroneousdetection.

Second Comparative Example

FIG. 13 illustrates the layout of a virtual receiving array according tothe second comparative example. For the sake of comparison, the numberof elements in the receiving antenna according to the second comparativeexample is equal to the number of elements in the receiving antennaelements Rx #1 through Rx #8 illustrated in FIG. 10A, Further, thenumber of elements in the transmitting antenna according to the secondcomparative example is equal to the number of elements in thetransmitting antenna elements Tx #1 and Tx #2 illustrated in FIG. 10B.

The eight elements of the receiving antenna are laid out equidistantlyin the first axis direction as illustrated in FIG. 11A, to maximize theaperture length of the virtual receiving array in the first axisdirection. Further, the two elements of the transmitting antenna arelaid out at a spacing of 8×d_(H) in the first axis direction, asillustrated in FIG. 10B. In this case, the layout of the virtualreceiving array is the layout illustrated in FIG. 13.

In a case where there are multiple targets at the same distance and thesame speed in a situation where the virtual receiving array having thelayout illustrated in FIG. 13 is being used, error will occur inestimation of the vertical direction component of the direction ofarrival, for example. On the other hand, a pair of virtual receivingarrays are arrayed in the second axis direction in the virtual receivingarray according to the first embodiment, so error in estimation of thevertical direction component is small even in a case where there aremultiple targets at the same distance and the same speed, so detectionprecision is improved.

An arrangement where the spacing of the transmitting antenna elements Tx#1 and Tx #2 of the transmission array antenna 108 in the first axisdirection is equal to the synthesized aperture length dR of thereceiving array antenna 202 is preferable, since the aperture length ofthe virtual receiving array in the first axis direction can be maximizedwhile arraying at least one pair of virtual receiving arrays in thesecond axis direction. For example, the synthesized aperture length dRof the receiving array antenna 202 illustrated in FIG. 10A, and thespacing of the transmitting antenna elements Tx #1 and Tx #2 of thetransmission array antenna 108 illustrated in FIG. 10B are both equal at8×d_(H). However, the spacing between the transmitting antenna elementsTx #1 and Tx #2 of the transmission array antenna 108 is not restrictedto this arrangement, and an arrangement may be made where the spacingbetween the transmitting antenna elements Tx #1 and Tx #2 is thesynthesized aperture length dR of the receiving array antenna 202 orless, and is an integer multiple of the first spacing d_(H). Having thespacing between the transmitting antenna elements Tx #1 and Tx #2 in thefirst axis direction to be narrower increases combinations with virtualreceiving arrays arrayed in the second axis direction, thereby improvingprecision in the vertical direction.

In the first layout example illustrated in FIG. 10A, the third spacingdu between the receiving antenna elements Rx #7 and Rx #8 is equal to2×d_(H). However, the size of the third spacing du is not restricted tothis. For example, enlarging the third spacing du and enlarging theaperture length of the virtual receiving array enables the main lobe ofthe formed beam to be narrowed, thereby improving resolution. Also, forexample, narrowing the third spacing du and narrowing the aperturelength of the virtual receiving array enables the sidelobe level to bereduced.

Second Layout Example

In the first layout example illustrated in FIG. 10A, the end portion ofthe receiving array antenna 202 is laid out non-equidistantly, so thatthe spacing between the receiving antenna elements Rx #7 and Rx #8 isdifferent from the spacings between the receiving antenna elements Rx #1through Rx #7. However; the antenna layout according to the firstembodiment is not restricted to the first layout example.

FIG. 14A illustrates a second layout example of receiving antennaelements Rx #1 through Rx #8 of the receiving array antenna 202according to a second layout example of the first embodiment. In thesecond layout example illustrated in FIG. 14A, the inner side of thereceiving array antenna 202 is laid out non-equidistantly, such that thespacing between the receiving antenna elements Rx #5 and Rx #6 isdifferent form the spacings between the receiving antenna elements Rx #1through Rx #5 and receiving antenna elements Rx #6 through Rx #8.

FIG. 14B illustrates the layout of a virtual receiving array accordingto the second layout example. As illustrated in FIG. 14B, the pair ofvirtual antenna elements VA #8 and VA #9 are laid out adjacent at aspacing of d_(V) in the second axis direction. It can also be seen fromFIG. 14B that the aperture length of the virtual receiving array in thefirst axis direction is equal to 16×d_(H).

FIG. 15 is a cross sectional view of two-dimensional beams according tothe first layout example and the second layout example at 0 degrees inthe second axis direction, taken along the first axis direction. It canbe seen from FIG. 15 that changing the receiving antenna elementssituated non-equidistantly in the receiving array antenna 202 changesthe sidelobe level. On the other hand, the aperture length of thevirtual receiving arrays are both equal to 16×d_(H) even if thereceiving antenna elements situated non-equidistantly in the receivingarray antenna 202 are changed, and there also is hardly any change inthe main lobe width.

Note that the layouts of the transmission array antenna 108 andreceiving array antenna 202 are not restricted to the above-describedlayouts. For example, even if the layouts of the transmission arrayantenna 108 and the receiving array antenna 202 are interchanged, avirtual receiving array that is the same as before interchanging isobtained, and properties the same as before interchange are obtained.Further, the layouts of the transmission array antenna 108 and receivingarray antenna 202 may be horizontally inverted and/or verticallyinverted.

In the first embodiment of the present disclosure, the radar device 10has the radar transmitting unit 100 that transmits radar signals fromthe transmission array antenna 108, and the radar receiving unit 200that receives returning wave signals of radar signals reflected at atarget from the receiving array antenna 202. Further, the layoutsillustrated in FIG. 9, 10A, 10B, or 14A are employed for thetransmitting antenna elements Tx #1 through Tx #Nt of the transmissionarray antenna 108 and the receiving antenna elements Rx #1 through Rx#Na of the receiving array antenna 202.

According to the first embodiment of the present disclosure, a virtualreceiving array that suppresses deterioration of resolution in thehorizontal direction and has angular estimation capabilities in thevertical direction can be configured, and a radar device capable ofthree-dimensional measurement can be configured, that is highly precisein the horizontal direction and that performs angular estimation in thevertical direction. Further, according to the first embodiment of thepresent disclosure, a MIMO radar device capable of three-dimensionalmeasurement, with additional angular estimation capabilities in thesecond axis direction, can be configured without deterioration ofangular separation capabilities in the horizontal direction as comparedwith a MIMO radar device performing one-dimensional beam scanning.

Second Embodiment

A second embodiment, where the layout of transmitting antenna elementsTx #1 through Tx #Nt of the transmission array antenna 108 differs fromthe antenna layout in the first embodiment, will be described below.Note that the configuration of the radar device 10 according to thesecond embodiment is generally the same as the configuration of theradar device 10 according to the first embodiment illustrated in FIG. 1except for the layout of the transmitting antenna elements Tx #1 throughTx #Nt of the transmission array antenna 108, so description of theconfiguration of the radar device 10 will be omitted.

Antenna Layout in Radar Device 10

Third Layout Example

FIG. 16A illustrates a third layout example of the transmitting antennaelements Tx #1 through Tx #Nt of the transmission array antenna 108according to the second embodiment. In FIG. 16A, the sum Nt of thetransmitting antenna elements #1 through #Nt is equal to four. Thetransmitting antenna elements #1 through #4 are laid out at the secondspacing d_(V) in the second axis direction, and also shifted by thefirst spacing d_(H) in the first axis direction every other one. Thefirst spacing d_(H) and the second spacing d_(V) respectively are, forexample, equal to 0.5 wavelengths and 0.6 wavelengths.

Each of the antenna elements of the transmission array antenna 108 havethe phase center at points (blank circles) illustrated in FIG. 16A, andare configured with the aperture length widened to a level whereantennas do not interfere with each other in the second axis direction,and the beam width is narrowed in the vertical direction. In a case ofusing the transmission array antenna 108 for a near-range wide-angleobservation application, the field of view (FoV) may be set wide in thehorizontal direction and the vertical direction. For example, the FoV isaround 80 degrees in the horizontal direction and around 30 degrees inthe vertical direction.

The layout of the multiple receiving antenna elements #1 through #Na ofthe receiving array antenna 202 according to the third layout example ofthe second embodiment is the same layout as the layout illustrated inFIG. 10A. Antenna elements with the aperture length widened in thesecond axis direction so that the beam width in the vertical directionis around 30 degrees, which is the field of view, with the pointsillustrated in FIG. 10A as the phase centers, are used for each of theantenna elements in the receiving array antenna 202.

The multiple transmitting antenna elements #1 through #Nt and themultiple receiving antenna elements #1 through #Na may be antennashaving a wide beam width so as to be able to observe a wide angle in thehorizontal direction. The antenna elements may be configured usingsub-array antennas, an array weight may be applied to the sub-arrayantenna to suppress sidelobes.

Dummy antenna elements may be disposed for the receiving antennaelements Rx #1 through Rx #8 laid out non-equidistantly as illustratedin FIG. 10A. For example, dummy antenna elements may be disposed innon-equidistant regions such as between the receiving antenna element Rx#7 and the receiving antenna element Rx #8, a region to the left side ofthe receiving antenna element Rx #1, or a region to the right side ofthe receiving antenna element Rx #8. Disposing dummy antenna elementsyields the advantages of uniform effects of electric properties such asantenna radiation, impedance matching, and isolation.

FIG. 16B illustrates the layout of a virtual receiving array accordingto the third layout example. The fact that the layout of the receivingantenna elements Rx #1 through Rx #8 and transmitting antenna elementsTx #1 through Tx #4 according to the second embodiment is an antennalayout that has high resolution in the horizontal direction and thevertical direction will be demonstrated below with by way of a thirdcomparative example where the receiving antenna elements are laid outequidistantly.

Third Comparative Example

For the sake of comparison, the number Na of elements in the receivingantenna elements Rx #1 through Rx #8 according to the third comparativeexample is equal to the number of elements in the receiving antennaelements Rx #1 through Rx #8 illustrated in FIG. 10A which is eight.Also, the number Nx of elements in the transmitting antenna elements Tx#1 through Tx #Nx according to the third comparative example is equal tothe number of elements in the transmitting antenna elements Tx #1through Tx #4 illustrated in FIG. 16A, which is four.

In the third comparative example, the receiving antenna elements Rx #1through Rx #8 are laid out equidistantly as illustrated in FIG. 11A. Thetransmitting antenna elements Tx #1 through Tx #4 are laid out in thesame way as illustrated in FIG. 16A.

FIG. 17 illustrates the layout of a virtual receiving array according tothe third comparative example. The aperture length in the first axisdirection of the virtual receiving array according to the thirdcomparative example is 8×d_(H), as illustrated in FIG. 17. This aperturelength is smaller than 9×d_(H), which is the aperture length in thefirst axis direction of the virtual receiving array according to thesecond embodiment illustrated in FIG. 16B.

FIG. 18 is a cross sectional view of two-dimensional beams according tothe third layout example and the third comparative example at 0 degreesin the second axis direction, taken along the first axis direction. Thebeam according to the second embodiment has lower adjacent sidelobes ascompared to the beam according to the third comparative example, asillustrated in FIG. 18. That is to say, the antenna layout according tothe second embodiment reduces the probability of erroneous detection ascompared to the antenna layout according to the third comparativeexample. Also, the beam width according to the second embodiment issmaller than the beam width according to the third comparative example.That is to say, the antenna layout according to the second embodimentyields higher resolution as compared to the antenna layout according tothe third comparative example.

In the second embodiment, the third spacing du between the receivingantenna elements Rx #7 and Rx #8 is equal to 2×d_(H) in the layoutillustrated in FIG. 10a , in the same way as in the first embodiment.However, the size of the third spacing du is not restricted to this. Forexample, enlarging the third spacing du and enlarging the aperturelength of the virtual receiving array enables the main lobe of theformed beam to be narrowed, thereby improving resolution. Also, forexample, narrowing the third spacing du and narrowing the aperturelength of the virtual receiving array enables the sidelobe level to bereduced.

In the second embodiment, changing the receiving antennas situatednon-equidistantly in the receiving array antenna 202 changes thesidelobe level, in the same way as in the first embodiment. On the otherhand, the aperture length of the virtual receiving array is unchangedeven if the receiving antenna elements situated non-equidistantly in thereceiving array antenna 202 are changed, and there also is hardly anychange in the main lobe width.

Fourth Layout Example

FIG. 19 is a diagram illustrating a fourth layout example of thetransmitting antenna elements Tx #1 through Tx #4 of the transmissionarray antenna 108 according to the second embodiment. In the secondembodiment, even in a case where the transmitting antenna elements Tx #2and Tx #4 are not shifted in the first axis direction by equidistantlylaid out at the second spacing d_(V) in the second axis direction, as inthe fourth layout example illustrated in FIG. 19, the same advantages asthose of the third layout example illustrated in FIG. 16A can beobtained.

The layouts of the transmission array antenna 108 and receiving arrayantenna 202 in the second embodiment are not restricted to theabove-described layouts, in the same way as in the first embodiment. Forexample, even if the layouts of the transmission array antenna 108 andthe receiving array antenna 202 are interchanged, a virtual receivingarray that is the same as before interchanging is obtained, andproperties the same as before interchange are obtained. Further, thelayouts of the transmission array antenna 108 and receiving arrayantenna 202 may be horizontally inverted and/or vertically inverted.

In the second embodiment of the present disclosure, the radar device 10has the radar transmitting unit 100 that transmits radar signals fromthe transmission array antenna 108, and the radar receiving unit 200that receives returning wave signals of radar signals reflected at atarget from the receiving array antenna 202. Further, the layoutsillustrated in FIG. 16 or 19, for example, are employed for thetransmitting antenna elements Tx #1 through Tx #Nt of the transmissionarray antenna 108 and the receiving antenna elements Rx #1 through Rx#Na of the receiving array antenna 202.

According to the second embodiment of the present disclosure, a MIMOradar device capable of three-dimensional measurement, which is capableof suppressing horizontal direction sidelobes of beams formed by thevirtual receiving array, and further capable of improving resolution inthe horizontal direction, with reduced probability of erroneousdetection, can be configured.

Third Embodiment

A third embodiment, where the layout of transmitting antenna elements Tx#1 through Tx #Nt of the transmission array antenna 108 is switched inusage, will be described below. FIG. 20 is a block diagram illustratingan example of the configuration of a radar receiving unit (radarreceiving circuit) 200 a according to the third embodiment.

A direction estimating unit (direction estimating circuit) 214 a of theradar receiving unit 200 a has functions of the direction estimatingunit 214 according to the first and second embodiments. Further, thedirection estimating unit 214 a inputs control signals from the controlunit 400, and switches operation modes of the radar device 10 based onthe control signals. Operation modes will be described layer withreference to FIG. 23. The components of the radar receiving unit 200 aother than the direction estimating unit 214 a are the same as thecomponents of the radar receiving unit 200 according to the first andsecond embodiments, so description will be omitted.

Antenna Layout in Radar Device 10

Hereinafter, the third embodiment will be described by way of an examplewhere the count alt of elements of the transmitting antenna elements Tx#1 through #Nt of the transmission array antenna 108 is equal to six,and the count Na of elements of the receiving antenna elements Rx #1through #Na of the receiving array antenna 202 is equal to eight.However, the counts of elements are not restricted to these numbers.

In the third embodiment, the receiving antenna elements Rx #1 through #8are partly laid out non-equidistantly, with a equidistant layout as thebasis, in the same way as in the first and second embodiments. Forexample, the layout of the receiving array antenna 202 is the same asthe layout illustrated in FIG. 10A.

FIG. 21 illustrates the fourth layout example of transmitting antennaelements Tx #1 through Tx #6 of the transmission array antenna 108according to the third embodiment. The transmitting antenna elements Tx#1 through Tx #6 include a first transmitting antenna group G1 and asecond transmitting antenna group G2.

The first transmitting antenna group G1 includes transmitting antennaelements Tx #1 and Tx #2, and the antenna layout thereof is the same asthe layout of transmitting antenna elements Tx #1 and Tx #2 according tothe first layout example of the first embodiment illustrated in FIG.10B. The first transmitting antenna group G1 is used for long-rangenarrow-angle observation usages, for example.

The second transmitting antenna group G2 includes the transmittingantenna elements Tx #3 through Tx #6, and the antenna layout thereof isthe same as the layout of the transmitting antenna elements Tx #1through Tx #Nt according to the third layout example of the secondembodiment illustrated in FIG. 16A. The second transmitting antennagroup G2 is used for short-range wide-angle observation usages, forexample. The transmitting antenna group being used is switched betweenthe first transmitting antenna group G1 and second transmitting antennagroup G2 in accordance with the observation usage.

The first transmitting antenna group G1 and second transmitting antennagroup G2 each independently configure a virtual receiving array. Thevirtual receiving array correlation vector illustrated in FIG. 10C isconfigured by the first transmitting antenna group G1 and the receivingarray antenna 202 illustrated in FIG. 10A. The virtual receiving arrayillustrated in FIG. 16B is configured by the second transmitting antennagroup G2 and the receiving array antenna 202 illustrated in FIG. 10A.

The first transmitting antenna group G1 and second transmitting antennagroup G2 illustrated in FIG. 21 may have the basic interval in the firstaxis direction in common, and may have a different spacing for the basicinterval in the second axis direction. For example, a basic spacingd_(H) 1 in the first axis direction for the first transmitting antennagroup G1, and a basic spacing d_(H) 2 in the first axis direction forthe second transmitting antenna group G2, may both be 0.5 wavelengths.Also for example, the basic spacing d_(V) 1 in the second axis directionfor the first transmitting antenna group G1 may be equal to 0.5wavelengths, and the basic spacing d_(V) 2 in the second axis directionfor the second transmitting antenna group G2 may be equal to 0.6wavelengths.

As described above, the first transmitting antenna group G1 and secondtransmitting antenna group G2 each independently configure a virtualreceiving array. Accordingly, the antenna elements of the firsttransmitting antenna group G1 and second transmitting antenna group G2may be freely laid out, including size, unless physically interfering.

FIG. 22 illustrates an example of the layout of the antenna elements ofthe transmission array antenna 108 according to the fourth layoutexample of the third embodiment. The overall installation area of thetransmission array antenna 108 can be reduced by laying out thetransmitting antenna elements of the second transmitting antenna groupG2 between the transmitting antenna elements Tx #1 and Tx #2 of thefirst transmitting antenna group G1, as illustrated in FIG. 22.

The configurations of the antenna elements of the first transmittingantenna group G1 and second transmitting antenna group G2 may beconfigurations each suited for the field of view (FoV). For example, theaperture length of the antenna elements of the first transmittingantenna group G1 is widened in both directions of the first axisdirection and second axis direction as illustrated in FIG. 22, to narrowthe beam width in both the horizontal direction and vertical direction.Also, for example, the aperture length of the antenna elements of thesecond transmitting antenna group G2 is widened to a degree where thereis no interference among the antenna elements as illustrated in FIG. 22,to obtain a radiation pattern of a beam that is relatively wide anglealong the vertical direction. The antenna elements may be configuredusing sub-array antennas, an array weight may be applied to thesub-array antenna to suppress sidelobes.

Dummy antenna elements may be disposed for the receiving antennaelements Rx #1 through Rx #8 laid out non-equidistantly as illustratedin FIG. 10A in the third embodiment, in the same way as in the firstembodiment. For example, dummy antenna elements may be disposed innon-equidistant regions such as between the receiving antenna element Rx#7 and the receiving antenna element Rx #8, a region to the left side ofthe receiving antenna element Rx #1, or a region to the right side ofthe receiving antenna element Rx #8. Disposing dummy antenna elementsyields the advantages of uniform effects of electric properties such asantenna radiation, impedance matching, and isolation, for example.

FIG. 23 illustrates an example of time division switching control of thefirst transmitting antenna group G1 and second transmitting antennagroup G2 according to the third embodiment. In a case where the radardevice 10 is a time division multiplexing MIMO radar, the radartransmitting unit 100 temporally switches antenna combinations to beused for time division multiplexed transmission, based on controlsignals from the control unit 400. In a long-range mode that is anoperation mode for long-range narrow-angle observation, the radartransmitting unit 100 uses the transmitting antenna elements Tx #1 andTx #2 of the first transmitting antenna group G1 for time divisionmultiplexed transmission, as illustrated in FIG. 23. Also, in ashort-range mode that is an operation mode for close-range wide-angleobservation, the radar transmitting unit 100 uses the transmittingantenna elements Tx #3 through Tx #6 of the second transmitting antennagroup G2 for time division multiplexed transmission.

Also, in a case of an operation mode where the long-range mode andshort-range mode are combined, the radar transmitting unit 100temporally switches the first transmitting antenna group G1 and secondtransmitting antenna group G2 to be used for time division multiplexedtransmission. For example, the radar transmitting unit 100 switches alltransmitting antenna elements Tx #1 through Tx #6 by time division. Forexample, in time slots dur1, dur2, dur7, and dur8, the transmittingantenna elements Tx #1 and Tx #2 of the first transmitting antenna groupG1 are used for time division multiplexed transmission, as illustratedin FIG. 23. Also, in time slots dur8 through dur6, dur9, and dur10, thetransmitting antenna elements Tx #3 through Tx #6 of the secondtransmitting antenna group G2 are used for time division multiplexedtransmission. Note that the order of using the transmitting antennaelements Tx #1 through Tx #6 is not restricted to the order illustratedin FIG. 23.

Note that in the third embodiment, the radar receiving unit in FIG. 20may be used, and the direction estimating unit 214 a may switch theoperation mode of the radar device 10 based on control signalsindicating the operation mode input from the control unit 400. Also, inthe third embodiment, the radar transmitting unit illustrated in FIG. 2may be used, and the radar transmission signal generating unit 101 mayswitch the operation mode of the radar device 10 based on controlsignals input from the control unit 400.

In one example, the radar transmitting unit 100 may transmit radarsignals that have different signal properties regarding transmissioncycle or transmission bandwidth in accordance with the operation mode,based on instruction information from the control unit 400. For example,in a case of operating in the close-range mode, the radar device 10 maytransmit radar signals over a relatively board band, in order to obtainhigher distance resolution. On the other hand, in a case of operating inthe long-range mode, radar signals may be transmitted at a relativelyshort cycle, in order to observe objects moving at higher speeds.

In a case where the radar device 10 is a MIMO radar that multiplexes bycode division or frequency division, the radar transmitting unit 100 mayswitch power supply to the first transmitting antenna group G1 andsecond transmitting antenna group G2 in accordance with the operationmode. Switching power supply selects the transmitting antenna group tobe used, and the operation mode is switched.

In the third embodiment of the present disclosure, the radar device 10includes the radar transmitting unit 100 that transmits radar signalsfrom the transmission array antenna 108, and the radar receiving unit200 that receives returning wave signals of radar signals reflected at atarget from the receiving array antenna 202. The radar device 10 furtherswitches the virtual receiving array to be used between the virtualreceiving arrays configured in the first embodiment and the secondembodiment, for example, in accordance with the operation mode.According to the third embodiment of the present disclosure, a MIMOradar capable of three-dimensional measurement can be configured thatobtains the advantages of the first embodiment and second embodiment inaccordance with the operation modes corresponding thereto.

Although various embodiments have been described above with reference tothe drawings, it is needless to say that the present disclosure is notrestricted to these examples. It is clear that one skilled in the artwill be able to reach various alterations and modifications within thescope of the Claims, and such should be understood to belong to thetechnical scope of the present disclosure as a matter of course. Variouscomponents in the above-described embodiments may be optionally combinedwithout departing from the essence of the disclosure.

Although examples of configuring the present disclosure using hardwarehave been described in the above-described embodiments, the presentdisclosure may be realized by software in cooperation with hardware aswell.

The functional blocks used in the description of the above-describedembodiments typically are realized as large-scale integration (LSI) thatis an integrated circuit. An integrated circuit may control thefunctional blocks used in the description of the above-describedembodiments, and have input and output. These may be individually formedinto one chip, or part or all may be included in one chip. Also, whiledescription has been made regarding an LSI, there are different namessuch as integrated circuit (IC), system LSI, super LSI, and ultra LSI,depending on the degree of integration.

The circuit integration technique is not restricted to LSIs, anddedicated circuits or general-purpose processors may be used to realizethe same. A field programmable gate array (FPGA) which can be programmedafter manufacturing the LSI, or a reconfigurable processor where circuitcell connections and settings within the LSI can be reconfigured, may beused.

Further, in the event of the advent of an integrated circuit technologywhich would replace LSIs by advance of semiconductor technology or aseparate technology derived therefrom, such a technology may be used forintegration of the functional blocks, as a matter of course. Applicationof biotechnology, for example, is a possibility.

Summarization of Embodiments

The radar device according to the present disclosure includes a radartransmitting circuit that transmits radar signals from a transmissionarray antenna, and a radar receiving circuit that receives returningwave signals, where the radar signals have been reflected at a target,from a receiving array antenna. One of the transmitting array antennaand the receiving array antenna includes a plurality of first antennasof which phase centers are laid out along a first axis direction. Theother of the transmitting array antenna and the receiving array antennaincludes a plurality of second antennas of which phase centers are laidout at a second spacing along a second axis direction that is differentfrom the first axis direction. The plurality of first antennas include aplurality of antennas of which the phase centers are laid out at a firstspacing, and a plurality of antennas of which the phase centers are laidout at a third spacing that is different from the first spacing.

In the radar device according to the present disclosure, the spacing ofphase centers of adjacent antennas of the plurality of first antennas isequal to the first spacing excluding at least one spacing, and at leastone spacing is equal to the third spacing.

In the radar device according to the present disclosure, the thirdspacing is equal to an integer multiple of the first spacing.

In the radar device according to the present disclosure, the thirdspacing is equal to twice the first spacing.

In the radar device according to the present disclosure, the at leastone spacing is one spacing.

In the radar device according to the present disclosure, the phasecenters of the plurality of second antennas are laid out at spacingsequal to a synthesized aperture length of phase center of the pluralityof first antennas, along the first axis direction.

In the radar device according to the present disclosure, the count ofthe second antenna elements is equal to two.

In the radar device according to the present disclosure, the length inthe first axis direction of a range where the phase centers of theplurality of second antennas are laid out is equal to or less than asynthesized aperture length of phase center of the plurality of firstantennas, and the plurality of second antennas are laid out at a spacingthat is an integer multiple of the second spacing in the second axisdirection.

In the radar device according to the present disclosure, the phasecenters of the plurality of second antennas are laid out at the secondspacing in the second axis direction.

In the radar device according to the present disclosure, the pluralityof first antennas have a first antenna group and a second antenna groupconfigured of different virtual receiving arrays, and the radartransmitting circuit and the radar receiving circuit each performtransmission of the radar signals and reception of the returning wavesignals by switching between the first antenna group and the secondantenna group.

The present disclosure is suitable as a radar device that detects over awide-angle range, and can be installed in vehicles, for example.

What is claimed is:
 1. A radar device, comprising: a radar transmittingcircuit that transmits radar signals from a transmitting array antenna;and a radar receiving circuit that receives returning wave signals,where the radar signals have been reflected at a target, from areceiving array antenna, wherein one of the transmitting array antennaor the receiving array antenna includes a plurality of first antennas ofwhich phase centers are laid out along a first axis direction, whereinthe other of the transmitting array antenna or the receiving arrayantenna includes a plurality of second antennas of which phase centersare laid out at a second spacing along a second axis direction that isdifferent from the first axis direction, wherein the plurality of firstantennas include a plurality of antennas of which phase centers are laidout at a first spacing, and a plurality of antennas of which phasecenters are laid out at a third spacing that is different from the firstspacing, and wherein two or more spacing of phase centers of adjacentantennas of the plurality of first antennas is equal to the firstspacing, and at least one spacing of phase centers of adjacent antennasof the plurality of first antennas is equal to the third spacing.
 2. Theradar device according to claim 1, wherein the third spacing is equal toan integer multiple of the first spacing.
 3. The radar device accordingto claim 2, wherein the third spacing is equal to twice the firstspacing.
 4. The radar device according to claim 1, wherein the phasecenters of the plurality of second antennas are laid out at spacingsequal to a synthesized aperture length of phase center of the pluralityof first antennas, along the first axis direction.
 5. The radar deviceaccording to claim 4, wherein a count of the plurality of secondantennas is equal to two.
 6. The radar device according to claim 1,wherein a length in the first axis direction of a range where the phasecenters of the plurality of second antennas are laid out is equal to orless than a synthesized aperture length of phase center of the pluralityof first antennas, and the plurality of second antennas are laid out ata spacing that is an integer multiple of the second spacing in thesecond axis direction.
 7. The radar device according to claim 6, whereinthe phase centers of the plurality of second antennas are laid out atthe second spacing in the second axis direction.
 8. The radar deviceaccording to claim 1, wherein the plurality of second antennas have afirst antenna group and a second antenna group configured of differentvirtual receiving arrays, and wherein the radar transmitting circuit andthe radar receiving circuit each perform transmission of the radarsignals and reception of the returning wave signals by switching betweenthe first antenna group and the second antenna group.
 9. The radardevice according to claim 1, wherein phase centers of at least threeconsecutive antennas of the plurality of first antennas are spaced fromeach other by the first spacing.